Feed system

ABSTRACT

A FEED SYSTEM SUITABLE FOR SATELITE COMMUNICATION EARTH STATION ANTENNAS AND WHICH INCLUDES A THREE PORT POLARIZER ASSEMBLY, MULTIMODE HORN ASSEMBLY AND TRACKING DIFFERENCE MODE CIRCUITRY IS DESCRIBED. THE THREE PORT POLARIZER ASSEMBLY IS COUPLED TO THE HORN ASSEMBLY FOR PROVIDING FULL POLARIZATION MODE FLEXIBILITY FOR THE SUM MODES. THE MULTIMODE HORN ASSEMBLY PROVIDES EXCITATION AND CONTROL OF BOTH THE SUM AND DIFFERENCE MODES AND INHIBITS THE UNDERSIRABLE MODES. THE TRACKING DIFFERENCE MODE CIRCUITRY IS COUPLED TO THE MULTIMODE HORN TO PROCESS DIFFERENCE MODES AND PROVIDE THE AZIMUTH AND ELEVATION INFORMATION FOR MONOPULSE TRACKING.

P. FOLD ES FEED SYSTEM Feb. 2, 197i A TORNA Y Feb. 2, 1971 P. FOLDES3,560,976

FEED SYSTEM Filed Aug. 2l, 1968 4 Sheets-SheetI 2 ab( f/ TO QECEIVELTERMINAL.

/ -f To BLEvnT/ou maw :um bl INVENTOR PETEIL FoLDt-:5

BY M) Mm ATTO/WIEV P. FOLDES FEED SYSTEM Feb. 2, 1971 4 Sheets-Sheet 3Filed Aug. 21, 1968 Feb. 2, 1971 P, FOLDES 3,560,976

FEED SYSTEM Filed Aug. 21, 1968 4 Sheets-Sheet 4 INVENTOH PETEL FoLoes ybel/)w06 .United States Patent Oce Patented Feb. 2, 1971 3,560,976 FEEDSYSTEM Peter Foldes, Montreal, Quebec, Canada, assignor to RCACorporation, a corporation of Delaware Filed Aug. 21, 1968, Ser. No.754,373 Int. Cl. H04b 7/10; H01q 13/02, 15/24 U.S. Cl. 343-100 17 ClaimsABSTRACT OF THE DISCLOSURE A feed system suitable for satellitecommunication earth station antennas and which includes a three portpolarizer assembly, multimode horn assembly and tracking difference modecircuitry is described. The three port polarizer assembly is coupled tothe horn assembly for providing full polarization mode fiexibility forthe sum modes. The multimode horn assembly provides excitation andcontrol of both the sum and difference modes and inhibits theundesirable modes. The tracking difference mode circuitry is coupled tothe multimode horn to process the difference modes and provide theazimuth and elevation information for monopulse tracking.

BACKGROUND OF THE INVENTION This invention relates to microwavecommunication systems and more particularly to a wideband multimodemonopulse antenna feed system with high gain to noise temperature ratiowhich provides the simultaneous functions of receiving, transmitting andautomatic tracking, wherein the first two are accomplished by the use ofsum signals and the third is accomplished by using difference modesignals.

Feed systems capable of generating and receiving microwave energy in aplurality of modes have been developed and are known as multimode feedsystems. 'Such multimode feed systems are often used in monopulsetracking wherein the energy transmitted and received by the feed systemsis combined in such a manner that sum and difference radiation patternsare produced during transmission and/or reception. These patterns areanalyzed to determine the position of the object which may be either anaircraft, a missile, or a satellite or to provide automatic tracking ofthat aircraft, missile, or satellite. Monopulse tracking systems arediscussed in Introduction to Radar Systems by Merrill L. Skolnick,published 1962 by Mc- Graw-Hill Book Co. and Introduction to Monopulseby D. R. Rhodes, published in 1959 by McGraw-Hill Book Co.

The typical tracking feed system may include several horns or apertures.When only a small number of horns, such as in the four horn system, areused, the radiation patterns have undesirable characteristics whichlower the efciency of the system and have increased noise level. Someattempts have been made to produce an efficient low noise multimode feedsystem with a single aperture. The prior art single aperture devicesalthough operative have a relatively low gain to noise temperature ratiowhen they are used as feed systems for reflector type antennas and whenoperated over a wide range of frequencies. In addition, the prior artmultimode feed systems do not have full polarization mode exibility ineither the sum (communication) channel or the difference (tracking)channel.

Although many feed systems have been built in the past in which one ormore of the three functions of transmit, receive and tracking arehandled independently by separate parts of the feed, efficient systemswith operational simplicity are required with a single aperture in theregion of the focus of the primary or secondary reectors servingsimultaneously for all three functions over a relatively wide range offrequencies. Moreover, to obtain the best field distribution efficiency,an undivided radiating source aperture is desirable as it is achieved bya single horn. Also, since the polarization (linear or circular) mayvary from satellite to satellite, and the attitude of the linearpolarization of the satellite may vary, a full polarization modeflexibility is therefore desirable for optimum gain in the sum mode andfor tracking. Also, the receive and transmit polarization in practicalsystems must be maintained orthogonal (right vs. left circular ororthogonal linear polarization) in order to match the polarization modesof the feed to the employed polarization modes of the satellite.

It is an object of this invention to provide an improved multimode feedsystem operable over a relatively wide range of frequencies.

BRIEF DESCRIPTION OF INVENTION In accordance with the teachings of thepresent invention, a full polarization mode flexibility monopulsemultimode feed system is provided wherein the feed system includes amode coupler having a sum mode coupling port aperture and a separateplurality of difference mode port apertures symmetrically located aboutthe sum port. A sum or communication chanel propagation network iscoupled to the sum port and is responsive to the energy supplied to thesum port for providing the first plurality of sum modes. Likewise aseparate monopulse tracking network including monopulse bridge circuitryis coupled to the plurality of difference mode ports and is responsiveto the energy supplied to these ports for providing circularly polarizeddifference mode signals for the second plurality of modes. The novelfeed system also includes a horn and mode filter for controlling theamplitudes and phase relationships between the first and secondplurality of modes in a single aperture.

DESCRIPTION OF A PREFERRED EMBODIMENT A more detailed descriptionfollows in conjunction with the following drawings wherein:

FIG. 1 is a block diagram of the antenna feed system in accordance withone preferred embodiment of the present invention,

FIG. 2 is a sketch of the mode coupler used in accordance with thepresent invention,

FIG. 3 is a diagram of the wanted modes in the multimode horn,

FIG. 4 is a block diagram illustrating phase relationships for rightcircular polarization,

FIG. 5 is a perspective view of the mode filter,

FIG. 6 is a sketch of the side view of the mode lter,

FIG. 7 is a perspective view of the horn,

FIG. 8 is a perspective View of an orthogonal coupler,

FIG. 9 is a sketch of the side view of a 90 differential phase shifterin accordance with the present invention,

FIG. l0 is a sketch of the end view of the differential phase shifter ofF-IG. 9,

FIG. 11 is a sketch of the rotary polarizer assembly, and

FIG. 12 is a fragmentary view in section of the rotary joint taken inthe general direction of arrow A.

Referring to FIG. 1, a wideband tracking `feed system is providedincluding a multimode horn assembly 11, a three port polarizer assembly12 and tracking mode circuitry 13. The composite feed assembly -for asatellite communication network is required, for example, to cover awideband 5925 to 6425 rnHz. frequency band for transmit and a 3700 to4200 mHz. frequency band for receive. The three port polarizer assembly12 maintains propagation of the transmit or receive sum communicationsignals in orthogonal relationship while the monopulse differencesignals are processed at the tracking mode circuitry 13. The multimodehorn assembly 11 includes a mode coupler 16, a mode filter 17 and asingle aperture horn 18.

FIG. 2 is an end view of the mode coupler 16 looking from the modefilter 17 toward the coupler 16. The mode coupler 16 is constructed of alarge conically shaped port at the center of a square waveguide sectionand includes eight symmetrically located identical waveguide ports 26through 33. The ports 26 through 33 in corner pairs form four circularlypolarized exciting subapertures `with each port being orthogonallypolarized with respect to its corner paired port.

The center conically-shaped transducer port 25 is, for example, for theabove stated frequency band about 2.12 inches in diameter at thereceiver equipment end 34 of the mode coupler and increases to about 3inches in diameter at the antenna horn end 35. A step 36 is formed atthe end of the conical port 25 between the 3 inch diameter conical atthe end and a 4 inch square waveguide section at the mode lter end ofthe mode coupler 16. The step 36 is formed by the edge surface of thecoupler 1-6 extending in front of and in a plane forward of the planeincluding the openings for the ports 26 through 33 and the end opening35 for the conical coupler 25. The length of the conical, centertransducer port 25 in this example is about 4.5 inches. The step 36excites in the horn in addition to the basic TElo mode applied to themode coupler 16, the TE30 mode as well as a small amount of T1312, TM12,TE21 and TM21 modes. Refer to FIG. 3 for a diagram of the wanted TEN andTEBO modes.

The eight symmetrically located identical waveguide ports 26 through 33which are used for monopluse tracking terminate at the step 36 at whichthe TE30 mode excitation occurs. For an understanding of the monopulsetracking, the total step aperture 36 of the mode coupler 16 isconsidered for discussion as divided into four subapertures labeled a,b, c and d as shown in FIG. 4. Each corner of subaperture a, b, c and dof the mode coupler 16 has a respective horizontally polarized port 28,27, 32, 31 and a respective vertically polarized port 29, 26, 33 and 30.Also, the power in each subaperture is equal as provided by theshort-slot hybrids 44 through 47 as illustrated in the arrangement showniu FIG. 1 and 4.

To provide circular polarization in the feed Ifor monopulse trackingwhich is adaptable to receive or transmit right or left circularpolarization or linear polarization, the phase relationship between the-four subapertures a, b, c and d of the step aperture 36 should remainthe same. However, each aperture a, b, c and d should support the TEMmode (horizontal polarization) and the TEM, mode (vertical polarization)and these should be either delayed or advanced relative to` each otherto excite a circular polarized electromagnetic signal for eachsubaperture.

The monopulse tracking circuitry for providing the difference modes foreither linear polarization or right or left circular polarization isillustrated in FIGS. l and 4. The circuit consists of six short-slothybrids 44 through 47, 50 and 53 and two magic tee hybrids 51 and 52 andvarious bends in the waveguides and waveguide sections. The sizes anddimensions of these waveguides and `waveguide sections are optimized atthe receiver tracking frequency bands. Referring to FIGS. l, 2 and 4,each of the eight paired ports, 26 through 33, is coupled to itsorthogonal paired port at one of the yfour shortslot hybrids 44 through47. One terminal of each of the hybrids 44 through 47 is terminated atterminals 55 through 58 respectively, and the other terminal is combinedfor example, with the output of another one of the hybrids 44 through 47at hybrid 50 or 53. The proper ports of the hybrids 44 to 47 to beterminated depends upon the polarization, right or left circularpolarization. For right circular polarized signals, waveguide section 48is used with output terminals 14 and 15 coupled to hybrid 50 andwaveguide section 49 is used with output terminals 19 and 20 coupled tohybrid 53. For right circular polarized signals as shown in FIG. l,output terminals 1, 3, 5 and 7 of hybrids 44 through 47 are terminatedand are unused, and terminals 2, 4, 6 and 8 are used and are connectedto like numbered terminals in waveguide sections 48 and 49. For leftcircular polarized signals, terminals 2, 4, 6 and 8 are terminated andterminals 1, 3, 5 and 7 and waveguide sections 59 and 60 are used. Theinput terminals 1 and 3 of waveguide section 59 are connected to thelike numbered terminal in hybrids 44 and 45 and output terminals 14 and15 of waveguide section 59 are connected to hybrid 50. The inputterminals 5 and 7 of waveguide section 60 are connected to the likenumbered terminal of hybrids 46 and 47 and the output terminals 19 and20 of waveguide section 60 are connected to hybrid 53. Also, the rolesof the output terminals 61 and 62 change from elevation difference modesto azimuth difference modes and vise versa as indicated in FIG. l withrespect to the use of right or left circular polarized signals. Each ofthe four hybrids 44 through `47 which are coupled to the corner pairedports 26 through 33 provides a power division and provides a 90 relativephase shift between paired ports. The combined system of the foursubapertures each with the 90 relative phasing of the verticallypolarized T1310 mode and the horizontally polarized TEM mode with properphasing between subapertures a, b, c and d provides the verticallypolarized TE20 mode and horizontally polarized hybrid HE11=TEH+TM11modes in the mode filter 17, `which modes, when they are superimposedprovide the circularly polarized azimuth difference mode system. SeeFIG. 3 showing these wanted modes.

Referring to FIG. 4 and considering, for example, for right circularpolarized azimuth difference mode, the short-slot hybrids 50 and 53 andmagic tees 51 and 52 are arranged so that the phase relationships ateach port of the subapertures are as shown. The phase marked after thesymbol A in FIG. 4 shows the relative phases in the azimuth differencemode system. The properly phased (a and d in effective 0 phase, b and cin effective 180 phase) and vertically polarized TEM, modes in thesubapertures a, b, c and d excite the vertically polarized TE20 mode.The similarly phased and horizontally polarized TEM, modes in thesubapertures a, b, C and d wherein a and d effective phase is 0 and band c is effective 180 phase excite the horizontally polarized hybridmode. Since the physical polarization or the direction of the electricfield in subaperture b is opposite that of a and the physicalpolarization direction of subaperture c is opposite that of subapertured and the ports a. and d and c and d are physically located on opposedends of that field, the subapertures a and d in effect are cophased at180 and subapertures b and c in effect are cophased at phase. Thesimultaneous presence and the 90 phasing of the vertically polarizedT1320 mode and the horizontally polarized hybrid HE11=TE11+TM11 modeprovides the circularly polarized difference mode with zero radiatedfield in the vertical plane (horizontal or azimuth difference mode).

The sarne combined system of the four subapertures with an orthogonalset of proper phasings provides the vertically polarized hybridHE11=TMH+TE11 mode and horizontally polarized TEOZ mode in the modefilter, which modes, when they are superimposed provide the circularlypolarized elevation difference mode system. The phase relation betweenthe circularly polarized subapertures a, b, c and d is determined by theremaining four hybrids of the system which are two short-slot hybrids 50and 53 and the two magic tees 35 and 52.

Referring to FIG. 4 and considering, for example, the right circularlypolarized vertical difference mode, the hybrids 50 and 53 and magic tees35 and 52 are arranged so that the pshae relationships at each port ofthe subapertures a, b, c and d are as shown. The phase marked after thesymbol E or elevation shows the relative phases in elevation differencemode of the subapertures for the combined circularly polarizeddifference mode comprising the combined horizontally polarized TE02 modeand vertically polarized T Elfi-T M11 modes. See FIG. 3 showing thesewanted modes. In this example at subaperture a the horizontalpolarization E or elevation is 0 phase, at subapertures b and d thehorizontal polarization is 180 phase and at subaperture c the horizontalpolarization is 0 phase relative to the elevation terminal 61.

Since the physical polarization direction or the direction of theelectric field in subaperture b is opposite that of subaperture a andthe physical polarization direction of subaperture c is opposite that ofsubaperture d, therefore the effective electrical phase at subaperture bis 0 phase and at subaperture c is 180 phase. By means of subapertures aand b being 0 effective relative phase and subapertures c and d being180 effective phase between the subapertures for E or elevation system,the vertically polarized hybrid HEn mode and the horizontally polarizedT E02 modes are excited. The simultaneous presence y and 90 phasing ofthe vertically polarized HEM mode and the horizontally polarized TE02mode provides the circularly polarized elevation difference mode. Eachtriangle in the FIG. 4 indicates the phase between the hybrids, `whereasthe phase in the subapertures is indicated by the number following E orelevation and A or azimuth phase. The phase with respect to either theazimuth marked A or elevation marked E indicates the phase of thecorresponding terminal at the input of the eight ports of the foursubapertures a, b, c and d for right circular polarization. Operationfor left circular polarization is similar. The roles, however, of thefeed terminals 61 and 62 are interehanged when the polarization in thedifference mode changes from right circular polarization as shown inFIG. 4 to left circular polarization and thus therefore the azimuthdifference mode channel becomes the elevation difference mode.

The tracking mode circuitry at this point on in the direction of thereceiver and processing circuitry network for combining the differencemode signals conforms to standard monopulse bridge arrangements such asfound on p. 178 of Introduction of Radar Systems mentioned previously.

Referring back to FIG. 2, the center conical sum mode transducer port isspecially dirnensioned and formed having an inner ring 37 which is inthe example described above about 1/s inch thick and about 3 inches fromthe horn end of the port 25 in order to cut off the propagation of thegenerated difference modes (hybrid HE11=TE11+TM11 and TE20) back intothe sum system and thus avoid the deterioration of the difference modeaxial ratio. Each of the eight difference mode ports has tuning stubs 40therein for suppressing the unwanted TEN, TMm, TEM and TMm modes in thethroat of the horn and to reduce the coupling between the sum anddifference ports and between the various difference mode ports. Thehigher order difference modes generated by theeight element ring arrayafter being refiected and modified by the conical sum mode transition inthe mode coupler are finally filterd and phased by the mode filter 17and horn 18 to produce the desired difference mode aperture distributionand primary patterns.

With just the four subapertures a, b, c and d providing circularlypolarized waves radiating directly into space with the phasing betweenthe ports of the suby apertures, the feed system would degenerate to acircularly polarized four horn monopulse system. However, since the modecoupler radiates into a single horn instead of space, the system becomesa single horn multimode system which has a more efficient monopulseaperture distribution than a four horn system.

A wideband tracking feed system required for satellite communicationnetworks shall cover, for example, the 5925 to 6425 mHz. band fortransmit and the 3700 to 4200 mHz. band for receive operation. Theproper widet band phasing and control of the described modes for suchcases is provided by mode filter 17 and horn 18. The mode filter 17 isplaced between the mode coupler 16 and the horn 18 wherein thecommunication (sum) and tracking (difference) mode signals are combinedin a single aperture. Referring to FIGS. 5 and 6, the mode filter 17 isa square semi-exponential waveguide section having a ridge 41 on each ofthe four sides 42 to excite a wideband of frequencies. The mode filter17 shown in FIGS. 5 and 6 has approximately a half cosine functionprofile along the axis of propagation, a square cross section andsymmetrically located ridges 41 in its two main planes. This shape orany such appropriately tapered variation such as the half cosinefunction profile is arranged so that it results in a minimum squarecross section between the two terminating flanges of the mode filter 17.The half cosine function profile is only one example. Any one of manyappropriately tapered variations along the axis of propagation may beselected wherein the selection of the length l of the mode filter andthe rate of change in width w or shape of the profile (such as theamplitude of the abovedescribed cosine function) is arranged so thefrequency dependence of the phase between the TEZO, TEOZ and hybridHE11=TE11+TM11 difference modes on one hand and the frequency dependenceof the phase between the TEm and TESU sum modes on the other hand isvirtually eliminated, and thus a very wide band multimode operation canbe achieved. The square cross section of the mode filter 17 has aminimum width w of typically 2x and a minimum length l of 5A where A isthe geometrical means of the limits of the nearly one octave frequencyband represented by the lower receive and higher transmit frequenciesmentioned in the above example. By the dimensions of the filter being soselected, only the desired higher order sum and difference modes canpropagate through this filter section and the phase between these wantedmodes assures low side lobes in the E and H planes in the primarypattern and leaves the axial ratio for the circularly polarized sum anddifference modes unchanged. The exact dimensions of the above-describedmode filter for each particular case using the above-given rules iseasily achieved by anyone skilled in the microwave antenna art. Theoutput cross section of the mode filter is made large enough so thatonly a small amount of a differential phase shift takes place betweenthe modes in the multimode horn itself. The basic TEW and the higherorder T1330 sum modes generated by the step in the mode coupler 16 tothe four inch cross section as filtered and phased by the mode filterand horn finally produce the sum mode aperture distribution andpatterns. The mode lter 17 and the square semiexponential waveguidesection 18 follow in such a manner that the generated higher order modesarrive at the aperture of the horn in the proper phase. The term properphase means that the total aperture field of the horn is more or lessaxially symmetrically tapered in the complete, for example, 500 mMz.receive frequency band.

Referring to FIG. 7, the final shaping and the frequency stabilizing ofthe aperture field of the horn is achieved by the shape of the horn 18which has an increase in the fiare angle at the end of the horn. Thisincreased flare angle produces a relatively large quadric phase error inthe transmit frequency band and a smaller quadratic phase error in thereceive band thus bringing the transmitter band beamwidth even closer tothe receiver band beamwidth. The horn 18 also has fins 21 along thefiared portion on each of the four sides to approximate even more aGaussian shaped output pattern. The horn 18 also has a square waveguidesection 22 in each of the four corners to increase the speed of thetraveling waves in the extra fiare section at the low end of thefrequency band and doing so to provide a more uniform phase front at theaperture for low frequencies.

Thus far, we have concerned ourselves with the processing of thetracking mode signals and not the communication signal. Referring backto FIG. 1, the sum mode signals at the mode coupler 16 are fed to athree port polarizer section 12 through the conical center port 25. Thethree port polarizer includes an orthogonal coupler 71, a fixed 90differential phase shifter 73 and a rotating 90 differential phaseshifter 75.

Referring to FIG. 8, the orthogonal coupler 71 is constructed of asquare waveguide section 80 having a ridge 81 on the inside surface ofeach of the sides 82 to excite a wideband of frequencies. The coupler 71acts to combine the transmit signals from transmit terminal 70 and thereceive signals toward receive terminal 72 to a common square waveguidedimensioned and ridge loaded to propagate both signals. The transmittedsignals are, for example, in the TEN, mode and the receive signals are,for example, in the orthogonal TEM mode. Three internal irises 83 ornarrow metal strips are provided Within the orthogonal coupler 71 at thetransmit end of the coupler. These irises are placed orthogonally to theelectric field of the TEN mode of the transmitter. The TEW mode signalsfrom the transmitted are coupled toward the first fixed phase shifter 73with negligible reflection at the irises 83 and the orthogonal receivesignals from the first fixed phase shifter 73 in the TEM mode arereflected at irises 83 in the side arm of the orthogonal coupler 71 tothe receiver terminal 72. In this manner, these irises do not interferewith the transmitted wave, but act as a 90 elbow for the waves coming toreceiver terminal 72 through the side arm 84.

The first fixed phase shifter 73 illustrated in FIGS. 9 and 10 is asquare waveguide section having a ridge 86 on each of the four sides toexcite wideband of frequencies and is the next element in the directiontoward the horn 18 from the orthogonal coupler 71. This square waveguidesection is capable of propagating signals in both the TEN mode, forexample, for the transmitter signals, and the orthogonal TEM mode forthe receiver signal. The first phase shifter 73 has fifteen irises 88 inthis above-mentioned example in the lower right-hand corner of thewaveguide when viewing the waveguide in the direction of propagation andhas fifteen irises 88 in the upper left-hand corner of the waveguide.The series of corner irises 88 act in such a manner that an incomingwave with polarization parallel to the side wall of the square waveguidesuch as the TEN or T1201 mode is split into two equal power orthogonalcomponents and a nominally 90 differential phase shift takes placebetween the two components by the time these components are propagatedthrough the phase shifter 73. The end result of the power split anddifferential phase shift is that the unit converts the incoming linearlypolarized wave into a left circularly polarized wave or right circularlypolarized wave for a vertically or horizontally (transmit or receiverespectively) polarized wave respectively. For this combination ofpolarization, the irises 88 in the phase shifter 73 are in the upperleftand lower righthand corners, when one views the three port polarizerfrom the transmit input terminal. For the proper wide band operation ofthese irises y88 a post 90 has to be placed in the middle of each iris.These posts 90 resonate in the upper part of the frequency band forinstance at 6.2 gHz. in the above example and the irises 88 resonate inthe lower part of the frequency band for instance at 4 gHz. in the aboveexample. In this manner their combined impedance remains nearly constantas a function of frequency and wide band operation is obtained. Theresidual mismatch effects of these strips or irises are cornpensated bytuning screws 89 which can be used also to further optimize the axialratio produced by the phase shifter in the transmit and receivefrequency band in the above example.

Since the orthogonal coupler 73 assures orthogonality Ibetween thetransmitted and received waves, the corresponding received polarizationis right circularly polarized or left circularly polarized.

The next unit in the three port polarizer towards the antenna is therotating polarizer assembly 75, see FIG. 1l. The assembly consists of arotary joint 74 with a square to circular waveguide transition 96 andthen a circular to square waveguide transition 97, a rotating 90differential phase shifter 75 and a second rotary joint 76 with a squareto circular waveguide transition 105 and a short section of a circularwaveguide 106 coupled to the center port 25 of mode coupler 16.

Referring to FIGS. ll and 12, the first rotary joint 74 has a square tocircular waveguide transition section 96, a circular to square waveguidetransition 97 and a pair of coaxial rotary flanges 98 and 99l whereinthe gap 109 between the rotating flanges of the transitions is about0.025 inch. The coaxial fianges 98 and 99 are located coaxially withboth transitions and cover the gap between the transitions. Each of thecombined rotary joint coaxial flanges 98 and 99 has two grooves 100 and10.1 within the inner circumference of the rotary joint coaxial fiangecover wherein the distance from the end of the groove `100 or 101 in therotary joint to the end edge of the transitions is half a wavelength(M2) at the transmitter frequency for one groove and for the othergroove is half a wavelength (M2) at the receiver frequency. The grooves100 and 101 therefore act to provide two chokes, one resonant at thereceiver frequency to refiect back into the waveguide the receivedsignal and the other to reflect back the transmitted signal to thuslyreduce radiation losses at the rotary joint.

The first rotary joint 74 acts with respect to the circularly polarizedtransmitted wave as a double transformer with a square to circularwaveguide transition and a circular to square waveguide transitionconnected back to back. The output terminals of the first rotary joint74 have a square waveguide cross section supporting the properly phasedTEM, and TEM modes. The center of the rotary fiange 98 is circular. Thediameter of the circular section of the waveguide is about the same asthat of a side of a square so as to assure propagation of the circularTEH mode. The output of the first rotary joint 74 is coupled to arotating 90 differential phase shifter 75 which is electricallyidentical to the first fixed 9 differential phase shifter 73 in FIGS. 9and l0, having therein the 30 irises and posts with l5 of each in theappropriate two corners.

The rotating polarizer assembly accepts the circularly polarized wavesfrom the fixed phase shifter 73 transforms these waves back to linearpolarized waves at the outputs of the rotary joint 76. The attitude ofthe output polarization is now however an orientation function of therotating polarizer and the orthogonal relationship between the receiveand transmit band waves remains at all attitudes. The output of the 90differential phase shifter 75 toward the horn is fed to a second rotaryjoint 76 similarly built as shown and described in connection with FIG.l2 with the chokes resonating at the receive and transmit ybandsrespectively. The output of the second rotary joint 76 is applied to the2.125 inch diameter center circular waveguide port which is part of themode coupler 116. Since the output polarization is always parallel toone side of the square waveguide, the output terminal of the fixed 90polarizer and the position of the rotating polarizer determines theorientation of the linearly polarized transmitted waves. The rotatable90 polarizer may be rotated using a motor 110 and gears 111 and `112 toalign the sum mode circuitry with the linear polarization of thesatellite. The attitude of the linear polarization is a function of theangle between the two phase Shifters. The original orthogonalrelationship however between the transmit and receive frequency bandwaves remains in all the attitudes. By removing the second phase shifter75 and the two rotary joints 74, 76 from the system, right or leftcircularly polarized signal operation is obtained for use with thetransmission and reception of circular polarized signals at the horn.

What is claimed is: 1. A monopulse multimode feed system for propagatingenergy supplied thereto over a wide band of frequencies and comprising:

a mode coupler having a sum mode coupling port and a separate pluralityof difference mode ports,

first means coupled to said sum port and responsive to said energysupplied thereto for providing a first plurality of modes having atleast one state of polarization,

second means coupled to said separate plurality of difference mode portsand responsive to said energy supplied thereto for providing a secondplurality of modes which are circularly polarized,

and third means coupled to said mode coupler including au energypropagating horn having a single aperture for controlling the amplitudesand phase relationships over said wide band of frequencies between theenergy in said first and second pluralities of modes for providingradiation patterns Iwith low side lobes in the E and H planes in theprimary pattern and for maintaining the axial ratio for the circularlypolarized difference modes.

2. The monopulse feed system as claimed in claim 1 wherein said sum modecoupling port is conically shaped.

3. The monopulse feed system as claimed in claim 1 wherein said modecoupler has a square waveguide cross section at the end of the couplerin the direction of said horn.

4. The monopulse feed system as claimed in claim 3 wherein said modecoupler includes means at said sum mode coupling port and within saidsquare waveguide cross section whereby said first plurality of modesincluding the symmetrical TEN and TE modes and TEU and TM21 and TEM andTMm modes are excited into said third means.

5. The monopulse feed system as claimed in claim 4 wherein said separateplurality of difference mode ports are orthogonally paired and arelocated symmetrically about said sum mode port.

6. The monopulse feed system as claimed `in claim S 10 wherein saiddifference mode ports are eight in number and wherein a pair of saiddifference mode ports is orthogonally positioned at each of the fourcorners of said square waveguide cross section.

7. The monopulse feed system as claimed in claim 6 wherein said secondmeans includes four differential phase shifter and power splitters witha different one of said phase shifter and power splitters coupled toeach pair of said orthogonally paired waveguide ports to providecircular polarization in said feed.

8. The monopulse feed system as claimed in claim 7 wherein each of said90 differential phase shifted and power splitters is a short-slothybrid.

9. The monopulse feed system as claimed in claim 8 wherein said secondmeans includes means for properly phasing said paired waveguide portsrelative to each other to excite TE20, T1502, and hybrids T Ell-j-TMHmodes in the output of the mode coupler and to provide simultaneouspresence and 90 phasing of the vertically polarized TE20 mode andhorizontally polarized hybrid T Ell-f-TMH mode whereby a circularlypolarized difference mode with a null plane in one plane is obtained andto provide the simultaneous presence and 90 phasing of the horizontallypolarized TE02 mode and vertically polarized hybrid TEll-i-TMH modewhereby a circularly polarized difference mode with a null plane in aplane orthogonal to said one plane is obtained.

10. The monopulse feed system as claimed in claim 9 wherein said meansfor properly phasing said paired waveguide ports includes a fifth andsixth hybrid with the output of a first one of said hybrids combinedwith the output of a second one of said hybrids at said fifth hybrid andwith the output of a third one of said hybrids combined with the outputof a fourth one of said hybrids at said sixth hybrid.

`11. The monopulse system as claimed in claim 1 wherein said third meansincludes a mode filter having a single square aperture cross section andwherein the length and the change in width of said square aperture crosssection of said mode filter is determined so as to control the frequencydependency of the phasing of the TElo and TEBO modes over said wide bandof frequencies to assure low side lobes in the primary pattern and tocontrol the frequency dependency of the phasing of the T1520, TEM andhybrid HE11=TEH+TMH modes over said wide band of frequencies to maintainthe axial ratio for the difference modes.

12. The monopulse feed system as claimed in claim 11 wherein the minimumwidth is on the order of 2A and its length is on the order of 5k, whereA is the geometrical means of the limits of an operating band of saidwide band of frequencies.

13. The monopulse feed system as claimed in claim 12 wherein said squarewaveguide section has a half cosine function profile.

14. The combination as claimed in claim 1 wherein said horn is a squarepyra-midal horn having a flare angle close to the aperture of the hornthat increases substantially so as to produce a large quadratic phaseerror at the higher frequencies of said wide frequency band to bring thehigher frequency band beamwidth close to the lower frequency bandbeamwidth.

15. The combination as claimed in claim 6 wherein said conically shapedport has an inner iris ring in order to cut off the propagation of thegenerated difference mode back into the sum system.

16. The combination as claimed in claim 6 wherein each of the eightdifference mode ports has tuning stubs therein for suppressing theunwanted TE12, TM21 and TEzl-l-TMIZ modes and to reduce coupling betweenthe sum and difference ports and between the difference ports.

17. The combination as claimed in claim 1 wherein said first meansincludes:

a first 90 differential phase shifter responsive to linearly polarizedsignals applied thereto and converts the linearly polarized waves intoopposite sense cir- References Cited Cularly polarized waves, UNITEDSTATES PATENTS a second 90 differential phase shifter coupled between 3277 485 10/1966 Howard 343 16(SD)X the first phase shifter and the sumports responsive 311516 3/1967 Packard 343 100(.3) to said circularlypolarized waves present at its input 5 3,353,183 11/1967 Giger 34316(SD)UX to convert the circularly polarized waves present at its inputinto linearly polarized waves, and RICHARD A- PARI-EY, Primary Examinermeans for rotating one of said phase Shifters relative T H TUBBESING,Assistant Examiner to the other where the attitude of the linearpolarization is a function of the angle between the two posi- 10 US C1'XR' tions of the two phase Shifters. 343-16, 756, 772, 783, 786

